Baseband-derived RF digital predistortion

ABSTRACT

A baseband-derived RF predistortion system using a lookup table having coefficients extracted at baseband and then applied at RF by means of a vector modulator. The architecture combines the narrowband advantage of envelope predistortion with the accuracy of baseband predistortion, and including compensation for memory effects. A polynomial-based alternative is also described.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/330,451, filed on Dec. 8, 2008, which claims the benefit of U.S.Provisional Patent Application No. 61/012,416, filed on Dec. 7, 2007,the disclosures of which are hereby incorporated by reference in theirentirety.

FIELD OF THE INVENTION

The present invention relates to power amplifiers for wirelesscommunications systems, and more particularly relates to predistortionmethods for linearizing the output of such power amplifiers.

BACKGROUND OF THE INVENTION

Reliable mobile, or wireless, communication systems rely on clean andconsistent transmission from base-stations under widely and rapidlychanging conditions. Therefore, the radio frequency (RF) poweramplifiers (PA) found in the base stations of such wirelesscommunication systems have typically been the most critical and costlycomponent. This is derived from the stringent requirements on spectrumand power efficiency of these transmitters, even though they are drivenby wideband and highly varying signals. To meet the demandingspecifications for these amplifiers, a number of linearizationtechniques have been implemented. One such linearization technique,called digital baseband predistortion, has been successfully implementedusing digital signal processors. However, digital baseband predistortionhas a disadvantage in that it requires the entire transmit path to beseveral times wider than the signal bandwidth due to the predistortedinput. Therefore, this wideband transmit path demands a fastdigital-to-analog converter (DAC) and wideband filters. Moreover, as thebandwidth of the input signal gets wider, the bandwidth requirement ofthe baseband predistortion system gets much wider. In contrast, the mainadvantage of RF envelope digital predistortion is that the transmit pathdoesn't need to be wideband. But RF envelope digital predistortion hasthe disadvantage that it requires additional components, such as anenvelope detector and large RF delay lines, that create inaccuracy andloss, as well as increased cost and complexity. There has therefore beena need for a predistortion system that provides the desired precisionwithout unnecessary cost and complexity.

SUMMARY OF THE INVENTION

The present invention comprises a new architecture for a predistortionsystem that substantially removes the wideband requirements andpotential distortions caused by the additional components typicallyrequired in the prior art. Experimental results demonstrate that theproposed architecture achieves a reduction of adjacent channel powerratio (ACPR) comparable to conventional baseband predistortion. Theproposed architecture is suitable for the applications which requirewide bandwidth (i.e., >100 MHz).

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates in block diagram form an embodiment of thepredistortion system of the present invention.

FIG. 2 illustrates in block diagram form a test bench for establishingand verifying the operation of the system of the present invention.

FIG. 3 illustrates in graphical form the measured spectra for the delaydependence of the system wherein:

(a) represents the output of a power amplifier without predistortion;

(b) represents the output of a power amplifier with one sample advanced;

(c) represents the output of a power amplifier with one sample delayed;

(d) represents the output of a power amplifier with coarse delay match.

FIG. 4 illustrates an alternative embodiment to the lookup table shownin FIG. 1, using a polynomial calculation.

DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS

A block diagram of an embodiment of the proposed system is shown inFIG. 1. A predistortion function, F, is derived at baseband as shown inthe block 100 labeled Digital Baseband Processing, but applied to theoutput at RF. A vector modulator 105, also indicated as VM, is employedto generate the predistorted signal based on the predistortion function.A magnitude calculation block 110 indexes an input signal in order todetermine the proper correction coefficients at each instant either froma lookup table or a polynomial calculation. A digital delay component isable to compensate for the delay difference, τ_(d), between thepredistorting path and the main transmit path, including memory effects.This delay mismatch between two paths should be compensated using delaycalibration.

More particularly, an input signal is converted into I and Q components120 and 125 in a conventional manner (and therefore not shown). The Iand Q components are provided to a digital delay 130, and also providedto the magnitude calculation block 110 as well as an adaptationalgorithm block 135.

In an embodiment as shown in FIG. 1, the magnitude calculation block andthe adaptation algorithm block both provide inputs to a look-up table140, which has stored therein a database of correction coefficientsappropriate to the range of operation of the system. As noted above, theLUT 140 supplies the correction factor appropriate to each value of theinput to a pair of DAC's 145A-B. The outputs of the DAC's 145A-B arefiltered in a pair of low pass filters 150A-B, the outputs of which areprovided to the vector modulator 105.

In at least some embodiments, the LUT values are initially obtained viaa calibration routine, whereby the output signal from the poweramplifier is measured and the LUT coefficients are estimated so as tocompensate for any nonlinear distortion created by the power amplifier.In such an embodiment, the LUT coefficients can be stored in the memoryof either an FPGA or a DSP, and can be populated with a microprocessoror digital signal processor. In addition, the LUT coefficients can beupdated by, as just one example, feeding the output of the poweramplifier back to the baseband signal processor, where it is comparedwith the input signal, in response to which the lookup table value isupdated. The feedback block shown in FIG. 1 can also be implemented by,for example, down-converting the output of the power amplifier tobaseband. The baseband signal can then be compared with the input signaland the resulting error will be used to update the LUT coefficients. Asjust one alternative, the output signal from the power amplifier can bespectrally monitored, and the feedback signal will monitor theout-of-band distortion using a downconverter, bandpass filter and powerdetector. The output of the power detector can then be used to adjustthe LUT values or, if a polynomial approach is used, to adjust thepolynomial coefficients.

Referring to FIG. 4, an implementation of the invention using apolynomial approach can be better appreciated. Instead of using a lookuptable as in FIG. 1, the output of the magnitude calculation block 110 isprovided to a series of polynomials F₁₁, F_(1R), F₂₁, F_(2R), F_(N1),F_(NR) (each calculated in either the DSP or FPGA performing otherprocessing), and then summed as shown. The result of the summation isthen provided to the DAC's 145A-B, just as with the output of the lookuptable in FIG. 1. The remainder of the design is the same, and istherefore not repeated in FIG. 4. The polynomials can be expressed asF ₁ {z}=α ₁₁+α₁₂ Z+α ₁₃ Z ²+ . . . +α_(1N) Z ^(N-1)andF ₂ {z}=α ₁₁+α₁₂ Z+α ₁₃ Z ²+ . . . +α_(1N) Z ^(N-1)

As noted above, the polynomial coefficients are updated in the samemanner as the updates to the lookup table described in connection withFIG. 1. It will be appreciated by those skilled in the art that thepresent invention, whether implemented using a polynomial approach, orthe lookup table approach of FIG. 1, is able to compensate for memoryeffects in the power amplifier, thus providing substantially improvedlinearization over the prior art.

Referring still to FIG. 1, the output of the digital delay 130 issupplied to a quadrature modulator 155, the output of which is suppliedto a DAC 160. The output of the DAC 160, which takes the form shown inthe graph 160A, is provided to a low pass filter 165, where it ismodulated in mixer 170 with the signal flo as indicated at 175, and thenpassed through bandpass filter 180. The result is provided to the vectormodulator 105, which also receives the correction signals, includingdelay compensation, from the LPF's 150A-B. The vector modulator outputforms the envelope indicated in the graph 105A, and is provided to thepower amplifier 185, the output of which is represented by the graph185A. The output is also sampled at 190, and the sample is fed backalong feedback path 195 as another input to the adaptation algorithmlogic 135, to permit the output signal to be monitored to ensure, amongother things, that the values in the look-up table are updated ifappropriate.

In some embodiments, the DAC 160 will preferably have at least twicebandwidth of the signal to be converted.

Delay mismatch: To see delay mismatch effects with respect toperformance of the system, suppose the RF input, x(t), consists of twotones with a tone spacing (ω₂−ω₁). The predistortion function, F, withdelay mismatch, τ_(d), can be described asF(t−τ _(d))=a ₁ +a ₃ |X _(e)(t−τ _(d))|² =a ₁+½a ₃+½a ₃ cos [(ω₂−ω₁)t+ω₁τ_(d)]  (1)

where X_(e)(t) is the envelope of the input signal, a's are the complexcoefficients of the polynomials, and τ_(d) is the delay mismatch. It canbe seen from (1) that the predistortion function requires the samebandwidth of the frequency spacing in order to compensate up to thirdorder intermodulation distortions (IMD). The predistorted input RFsignal, X_(PD)(t), then can be expressed asX _(PD)(t)=x(t)F(t−τ _(d))  (2)

After substituting (1) into (2), expanding, and arranging it, it can besimply formulated asX _(PD)(t)=b ₁ S+b ₃ S _(U) _(—) _(IMD)<−(ω₂−ω₁)τ_(d) +b ₃ S _(L) _(—)_(IMD)<(ω₂−ω₁)τ_(d)  (3)

where b's are complex coefficients and S, S_(U) _(—) _(IMD), and S_(L)_(—) _(IMD) denotes two tone RF input signals, third order upper IMD,and lower IMD components, respectively, and < represents the angle inrelation to the next term. From (3), depending on τ_(d), the phase ofthe upper IMD components is decreased by (ω₂−ω₁) τ_(d) and the phase ofthe lower IMD parts increases by the same amount.

Experimental Results: A single carrier wideband code divisionmultiplexing access (WCDMA) signal with 10 dB peak-to-average powerratio (PAPR) is used in the test bench shown in FIG. 2 for the proposedstructure of the predistortion system. The test bench consists of twoelectronic signal generators (Agilent E4433B and E4438C), a vectormodulator (Analog Devices AD8341), a Doherty power amplifier with 300WATTS peak envelope power (PEP) and 61 dB gain, a vector signal analyzer(Agilent VSA89641A), and a personal computer with MATLAB and advanceddesign system (ADS). The baseband in-phase (I) and quadrature (Q)outputs on the rear panel of E4438C are connected into AD8341. The firstsource (E4433B) is considered as a master and its 10 MHz referenceoutput is used by the slave source (E4438C) as a clock reference (10 MHzinput). The RF input signal x(t) and the baseband derived signal orfunction F are synchronized based on the following procedures. A markeris placed at the beginning of the input signal file x(t) in the mastersource, so that a pulse is sent on the EVENT1 output every time thatthis marker is met. The EVENT1 output is connected to the patterntrigger input of the slave. In order to estimate delay difference,coarse delay calibration was performed based on delay measurementsbetween the main path for the RF input signal and the baseband path forthe predistortion function. The predistortion algorithm applied here isbased on memoryless fifth order polynomial using indirect learning.

FIG. 3 shows the measurements results for the digital predistortionsystem of the present invention. The system reduces the distortionsaround 15 dB as seen from curves (a) and (d) in FIG. 3, and theperformances with respect to delay dependence are represented in (b) and(c). With one sample (26 nsec) advanced and one sample delayedintentionally to investigate effects on delay, the system performancesare degraded by around 4 dB to 10 dB. This verifies that delay mismatchis detrimental to the proposed system performance like RF envelopedigital predistortion. However, using the system of the presentinvention, the delay can be substantially perfectly matched usingdigital delay, unlike RF envelope digital predistortion which utilizesanalog RF delay lines.

Having fully described an embodiment of the invention and variousalternatives, those skilled in the art will recognize, given theteachings herein, that numerous alternatives and equivalents exist whichdo not depart from the invention. It is therefore intended that theinvention not be limited by the foregoing description, but only by theappended claims.

1. A wideband predistortion system for use with RF power amplifiers, thesystem comprising: a baseband input operable to receive a basebandsignal; an RF output operable to provide a signal to an RF poweramplifier; a lookup table including a predistortion signal includingpredistortion components; a vector modulator coupled to the RF outputand operable to apply the predistortion signal to the RF output; and anadaptation algorithm unit operable to optimize the predistortioncomponents.
 2. The wideband predistortion system of claim 1 wherein thepredistortion signal includes predistortion coefficients based on adifference between the baseband signal and a feedback signal derivedfrom the RF output.
 3. The wideband predistortion system of claim 1wherein the vector modulator comprises a plurality of delayed and summedvector modulators.
 4. The wideband predistortion system of claim 1wherein the lookup table is indexed by a square of a magnitude of thebaseband signal.
 5. The wideband predistortion system of claim 1 furthercomprising a quadrature modulator coupled to the baseband input andoperable to receive the baseband signal and translate the basebandsignal to an RF frequency.
 6. The wideband predistortion system of claim1 wherein the vector modulator comprises a variable attenuator and avariable phase shifter.
 7. The wideband predistortion system of claim 1further comprising a digital delay block coupled to the adaptationalgorithm unit.
 8. The wideband predistortion system of claim 7 whereinthe digital delay block comprises an equalizer.
 9. A widebandpredistortion system for use with RF power amplifiers, the systemcomprising: a baseband input operable to receive a baseband signal; aquadrature modulator coupled to the baseband input and operable toreceive the baseband signal and translate the baseband signal to an RFfrequency; an RF output operable to provide a signal to an RF poweramplifier; a vector modulator coupled to the RF output and operable toapply the predistortion signal to the RF output, the vector modulatorcomprising a plurality of delayed and summed vector modulators; and alookup table coupled to the vector modulator and including apredistortion signal including predistortion components.
 10. Thewideband predistortion system of claim 9 wherein the lookup table isindexed by a square of a magnitude of the baseband signal.
 11. Thewideband predistortion system of claim 9 wherein the predistortioncoefficients are based on a difference between the baseband signal and afeedback signal derived from the RF output.
 12. The widebandpredistortion system of claim 9 wherein the vector modulator comprises avariable attenuator and a variable phase shifter.
 13. The widebandpredistortion system of claim 9 further comprising a digital delay blockcoupled to the quadrature modulator.
 14. The wideband predistortionsystem of claim 13 wherein the digital delay block comprises anequalizer.